1. Field of the Invention
This invention relates to AC electric drives and, in particular, to a method of induction motor control and an electric drive realizing this method.
This invention is of particular advantage in robotics and flexible manufacturing systems to control the torque, speed, and positioning of actuators of industrial robots, NC machine tools, and flexible manufacturing systems equipped with squirrel-cage induction motors, including assembly and welding robot systems.
2. Description of the Prior Art
The most challenging problem in developing automatic systems for rotor speed and position control in three-phase asynchronous or induction motors consists in providing control of the induction motor magnetic field and torque. The physical processes involved in their generation and control are determined by one and the same process of modifying the three-phase stator current and its parameters: phase, frequency, and magnitude.
In contrast to other types of electric motors, the energy characteristics of a variable-speed induction motor, regarded as an electromechanical energy converter, cannot be unambiguously described by the design characteristics of the induction motor or the physical properties of active parts of the stator and rotor, since it largely depends on the processes of induction motor magnetic field and torque control, which are, in turn, determined by the method and principles of control of the stator current parameters. Induction motor control is peculiar in that the same torque, speed, and power on the motor shaft can be obtained with substantially different magnitudes of the stator current and flux amplitudes, various saturation levels of the core, and, consequently, different redistribution and losses in power and heating in the active parts of the induction motor, e.g. in stator windings, stator steel, and shorted rotor bars.
In this connection, development of methods to control induction motor stator current and attempts to realize such methods within the framework of automatic speed control systems and induction motor servodrives responding to perform a specified change of position encounter serious difficulties when an integrated solution is sought to achieve contradictory targets involving performance and power requirements of an electric drive. Induction motor control has to be dealt with in terms of three basic targets:
1. Invariant control (independent of disturbance factors, including the magnitude and type of variations in the load moment and rotor speed) which is independent control of the induction motor torque, magnitude and angular position of the magnetic flux in accordance with the input control actions with arbitrary, including spasmodic, changes of this control input proportional to the desired induction motor torque.
2. Optimal control of stator current and magnetic flux is performed in steady-state conditions, when the rotor speed and induction motor torque are constant, to achieve the minimal stator current, minimal losses of power, and minimal heating of the stator winding insulation which is the most vulnerable active part of an induction motor; and, in dynamic conditions, to achieve the maximum torque with constraints being imposed on the stator current and voltage, and the temperature of the stator winding.
3. Linear control of the rotor speed and position should be performed, irrespective of the load moment, in conformity with the magnitude and type of control input changes over a broad range of rotor speed up to the zero speed and subject to the minimal control time and minimal steady-state and dynamic errors in the rotor speed and position in response to smooth or abrupt variations of the load moment. It should also permit holding zero speed for long periods with the load applied or when the load moment varies.
This package solution to the problem of induction motor control is required when such induction motors are employed as actuator motors of robots and flexible manufacturing systems where maximum efficiency is to be combined with positioning accuracy and repeatability irrespective of the load variations with minimal weight and restricted size of the electric actuating devices and electromechanical modules which are a combination of an electric motor, transducers, and mechanical transmission.
But induction motors are not used in robot technology precisely because the problems of control have not been yet adequately resolved. Obvious economic, operational, structural, and technological advantages of induction motors, as compared to DC motors and synchronous motors with permanent magnets, are defeated by shortcomings of known induction motor drives, such as inferior accuracy of rotor speed and position control, slower response time, lesser specific torque per unit weight and volume, and lesser torque-to-current ratio.
Even more efficient induction motor control systems based on frequency control are deficient in that the induction motor torque tends to fluctuate. This results in rotor speed pulsations, particularly when the rotor is loaded at infraslow speed which is close to zero speed. Changes of the torque become sluggish and non-linear, which results in slower response when controlling rotor speed and position, excessive correction of speed and position, which is inadmissible in robot technology. The induction motor is overheated due to excessive initial excitation currents flowing in the stator windings in the absence of the induction motor torque, which reduces the heating margin, considering frequent positioning cycles, adds to the load demand of the induction motor, and makes the actuating motor heavier and more bulky.
But if these disadvantages are eliminated, the use of induction motors in robotics looks extremely promising and advisable since it is at least ten times cheaper than DC and AC motors featuring high-efficiency permanent magnets made of rare earth metals. The specific power per unit weight of high-frequency three-phase induction motors with a rated frequency of 200 Hz and 400 Hz and rated synchronous speed of up to 12,000 rpm is at least twice as large. These advantages of induction motor drives become particularly evident in high-speed mechanical transmissions having a high gear ratio of 200-500, which is in line with the long-term prospects of development of robot technology. The use of precision induction motor drives having a wide speed range and no shaft-mounted sensors is particularly advisable for high-speed main drives of machine tools, high-speed electric spindles rated for at least 20,000 rpm, electrode wire feed drives of welding robots, fully automatic and semi-automatic arc welding machines, and plasma generators employed in flexible manufacturing technology.
Variable-speed induction motors rated for synchronous speed of 750-1,000 rpm are also economically advantageous when used in feed servodrives of NC machine tools in order to provide high accuracy and smoothness of motion at infralow rotor speeds and high dynamic rigidness of speed-to-torque characteristics when responding to abrupt variations of the load moment.
A unified closed-loop speed control induction motor for robot applications and flexible manufacturing systems should have a rotor speed control range of at least 10,000, the control range being defined as a ratio of the maximum rotor speed to its minimum speed, wherein the motion smoothness, as well as the linearity and rigidness of speed-to-torque characteristics are maintained. The rotor zero speed drift should be eliminated. At minimal speed, in the range of 0.1-1 rpm, the drop in speed caused by the load moment increasing to the motor rated torque should not exceed, respectively, 0.02-0-2 rpm; the amplitude of fluctuations of the minimal instantaneous speed should not go lower than 0.025-0.25 rpm; the minimal speed recovery time in case of a nominal load moment surge should not be lower than 0.01 sec. A variable-speed induction motor should be capable of executing the harmonic law of rotor speed control, prescribed by a control input, with a frequency not less than 100 Hz, while the rotor speed should correspond to not less than 0.707 of the amplitude of the harmonic control input with a phase lag not more than 90.degree.. In case of a step change in the speed control input, excessive correction of the actual speed of the rotor should not be more than 10%.
In a closed-loop induction motor servomechanism of a robot, position control should be performed without excessive correction, the positioning error being not more than one increment of a discrete position sensor. The dynamic error of the robot contouring control system should not exceed two discrete elements (increments).
In this case, the above quality characteristics of the rotor speed and position control should be achieved in conditions of the maximum shaft torque and power of the induction motor with certain constraints imposed on the stator current, stator voltage, and motor temperature.
Besides, it is advisable to make use of transducers commonly employed in robotics and flexible manufacturing systems. Rated power of actuating motors in robot technology commonly ranges from 10 W to 2.2. kW. It is, therefore, technologically difficult to insert field sensors, such as Hall-effect devices, into the air gap of an induction motor. It is also inadvisable from the point of view of operational reliability. In consequence, magnetic field should be controlled by methods whereby this magnetic field is measured indirectly, and this only further complicates the problem of induction motor control.
Known in the art are various methods of induction motor control by changing the stator current magnitude and frequency. Thus, for example, there has been proposed an induction motor control method (cf., for example, U.S. Pat. No. 3 824 437, Cl. HO 2P 5/40, 1974) comprising the steps of measuring the magnetic field in the air gap of an induction motor, measuring stator current, converting the measured stator current into two stator current quadrature components oriented in relation to the measured magnetic flux, adjusting one of the quadrature stator current components which is proportional to the desired amplitude of the rotor flux linkage so that it remains on a constant level assigned by a permanent control input corresponding to the desired constant amplitude of the rotor flux linkage, while the other quadrature stator current component is changed in proportion to a second control input whose magnitude is proportional to the desired torque of the induction motor.
The electric drive realizing this method comprises an induction motor whose stator windings are connected, via a stator current transducer, to outputs of an inverter, two Hall-effect devices being installed in the air gap of the induction motor and the outputs thereof being joined, via adders, with the outputs of the stator current transducer and coupled to inputs of a vector analyzer. Two outputs of the vector analyzer are connected to first and second inputs of a coordinate converter whose third input is connected to a unit for assigning a constant amplitude of the rotor flux linkage. The fourth input of the coordinate converter is connected to an output of a proportional-integral speed controller, while two outputs of the coordinate converter are connected to control inputs of two current regulators whose two other inputs are connected to the outputs of the current transducer. The outputs of the two current regulators are connected to inputs of a unit for converting a two-phase signal into a three-phase signal assigning the stator voltage, outputs of this converting unit being connected to control inputs of the inverter.
This control method is deficient in that the speed control range is not sufficiently broad, pulsations increase at infralow rotational speeds, control time in response to load disturbances is too long, the bandwidth of the speed control loop is narrow due to the use of sensed magnetic flux quantities as reference control inputs, these sensed quantities being subjected to discrete variations, especially at low rotational speed, because of minute serrations on the stator and rotor of the induction motor as a result of the machining process. Other disadvantages include inferior energy characteristics and a low ratio of the induction motor torque to the stator current, which is due to control based on the principle of constant amplitude of the rotor flux linkage. This results in substantial losses for magnetic field excitation with the torque being absent or insignificant. This is particularly true for low-power induction motors wherein the magnetizing current level is close to the stator rated current level
The electric drive realizing the disclosed method is deficient in that it is too complicated. Moreover, measurements of the magnetic flux by Hall-effect devices which should be fit into the serrated surface of the stator are often unreliable and the air gap between the stator chambers and the rotor surface is seriously reduced, which is particularly important for low-power induction motors whose air gap is already very narrow.
Also known in the art is a method of induction motor control (cf., for example, USSR Inventor's Certificate No. 193604, Cl. HO 2K, 1967), comprising the steps of controlling, phase-after-phase, instantaneous phase currents of the induction motor stator by comparing the commanded and sensed magnitudes of instantaneous stator phase currents, measuring stator current as a proportion of the quadrature sum of two stator current components, one component being constant and corresponding to the desired constant value of the magnetic flux, while the other current component is a variable which is changed proportionally to the input corresponding to the desired induction motor torque. Concurrently, the stator current frequency is changed in proportion with the sum of two frequencies, one frequency being the rotor speed, while the other is changed proportionally with the desired torque.
An electric drive realizing this method comprises an inverter whereto phase negative feedbacks are applied, which use instantaneous phase currents of the stator. The outputs of the inverter are connected to the stator windings of the induction motor, while three control inputs thereof are connected to the outputs of a synchro resolver whose shaft is connected to the shaft of the induction motor, two inputs of the synchro resolver are coupled to an electromechanical transducer of input control signals.
This method is deficient in that a change in the induction motor torque is oscillatory because the stator current frequency is changed as a sum of two frequencies. Another disadvantage is relatively poor performance due to control on the basis of the constant magnetic flux and oscillatory changes in the phase angle between stator current and magnetic flux.
The electric drive realizing this method is deficient in that the control procedure is too complicated, involving electromechanical conversion of control signals.
Also known in the art is a method of induction motor control (cf., for example, U.S. Pat. No. 4 418 308, Cl. HO 2P 5/34 1983) whereby stator windings of an induction motor are supplied with stator symmetrical three-phase voltage whose instantaneous phase values are changed by PDM (pulse-duration modulation) methods in accordance with the desired stator voltage amplitude and frequency, instantaneous stator phase currents and stator current amplitude are measured, and the stator voltage amplitude is controlled as an error between the desired and sensed amplitude of the stator current. After that, the rotor speed is measured, as well as the torque of the induction motor and the amplitude of the rotor flux linkage. The stator voltage frequency is controlled proportionally to the sum of two quantities, one quantity proportional to the sensed rotor speed, while the other quantity equal to the slip frequency is controlled as an error between the desired and sensed torque of the induction motor. The desired torque is produced as a function of an error between the desired and sensed rotor speed. The desired current amplitude of the stator is controlled as a function of the desired torque of the induction motor, taken as a square sum of two components of the stator current amplitude, one component being constant and corresponding to the desired constant amplitude of the rotor flux linkage, while the other is changed proportionally to the slip frequency. After that, the measured amplitude of the rotor flux linkage is compared with the constant value of the desired amplitude of the rotor flux linkage, which is assigned by a constant input. The desired amplitude of stator current is changed by the increment value of the stator current amplitude, which is controlled as an error between the amplitude of the rotor flux linkage assigned by the constant input and measured in the process of control.
The electric drive realizing this method comprises an induction motor whose stator windings are connected, via instantaneous phase current sensors, to outputs of a PDM voltage inverter. The rotor of the induction motor is connected to a tachometer generator. Three inductive pickups are installed in the induction motor. The outputs of the induction motor and outputs of instantaneous phase current sensors are connected to inputs of a processor calculating the rotor torque and flux amplitude, one such input being connected to a comparison unit of a torque regulator, while the other to a comparison unit of a rotor flux amplitude regulator. The output of the rotor flux amplitude regulator and the output of a stator current amplitude calculating unit are connected to two inputs of the comparison unit of the stator current amplitude regulator. The third input of the comparison unit of the stator current amplitude regulator is connected to the output of a stator current amplitude measuring unit whose inputs are connected to outputs of the instantaneous stator phase current sensors. The output of the stator current amplitude regulator is connected, via a correcting adder, to an amplitude setting input of the PDM voltage inverter. The second input of the correcting adder is connected, via a stator voltage amplitude correcting unit, to a frequency setting input of the PDM voltage inverter. The frequency setting input is connected to the output of an adder whose one input is connected to the output of the tachometer generator, while the other input is connected to an input of the stator current amplitude calculating unit and to the output of the torque regulator. The second input of the comparison unit of the torque regulator is connected to the output of the speed regulator whose input is connected to the output of the tachometer generator.
This method is deficient in that it provides narrow range and low accuracy of speed control at low rates and under load. Moreover, with the zero speed of the rotor the torque control response is sluggish, static and dynamic stability of the speed-torque characteristics is not sufficient due to oscillations, nonlinearity, and intertia of the torque control of the induction motor, which is caused by nonlinearity and ambiguity of the coupling of the stator voltage phase produced by the voltage inverter through the frequency control channel with instantaneous values of the induction motor torque and the amplitude of the rotor flux.
The electric drive realizing this method is deficient in that its power characteristics are poor due to appreciable heating of the stator winding insulation when the motor is idling at low torque. This is caused by the constant amplitude of the rotor flux and, at high dynamic ratings of the induction motor torque, by the low ratio of the maximum torque to the admissible amplitude of stator current.
Another disadvantage of this electric drive consists in that the control system is too complicated. Inductive pickups have to be installed inside the induction motor, induction motor torque has to be calculated on the basis of signals fed from instantaneous phase current sensors and inductive pickups; the instantaneous amplitude of stator current has to be calculated at a frequency close to zero. Moreover, it is not easy to adjust four regulators and a complex stator voltage amplitude correcting unit.
A general disadvantage of the disclosed method and electric drive consists in that special-purpose sensors and pickups are incorporated into the induction motor design. They are additional, as compared to other types of induction motor drives, components which are complicated, expensive, and not easy to manufacture and install. This feature seriously affects the basic advantages of an induction motor as an adjustable actuating motor for flexible manufacturing systems and robot technology, such as uncomplicated design, low cost, and adaptability to production processes.
Unstable dynamic and power characteristics of such induction motor drives during lengthy operational periods is another limitation connected with deviations of the actual electrical parameters of the induction motor from the adjusted parameters of the control system. Such deviations are not infrequent when the motor is heated during operation.
All these shortcomings prevent induction motor drives from being employed in robotics and flexible manufacturing systems instead of DC electric drives. More efficient control methods have to be devised to provide better dynamic characteristics, power performance, and accuracy of control. Such novel methods will have, for economic reasons, to do with a minimum number of standard sensors or pickups already widely used to control DC electric drives in many industrial systems.
It is advisable, therefore, to make use of indirect control methods involving some vectors characterizing the condition of an induction motor, e.g. rotor flux vector, and do without their direct measuring. In this case the well known field-oriented principle whereby a reference magnetic flux vector is measured and all control operations are referred to this measured vector has to be abandoned in favour of a more economically efficient and general principle of control synchronization, whereby the control input based on predetermined optimal laws is used to assign a control synchronization frequency which is found to be the most effective one in terms of utmost simplicity of the control system and its required accuracy. This control input is also used to produce optimal control processes in relation to a system of coordinates, which rotates with the synchronization frequency. Here the phase of the control synchronization signal is the reference phase and the phase of the controlled vector is produced in relation to this reference phase in accordance with optimal laws.
The prior art method which is the closest to the disclosed method of induction motor control (cf., for example, USSR Inventor's Certificate No. 1064411, Cl. HO2P 5/34, 1983) consists in that the rotor speed and induction motor torque are controlled in steady state conditions, and to this end the method comprises the steps of assigning a control input proportional to the desired amplitude of the rotor flux linkage, which is equal to the actual amplitude of the rotor flux linkage, producing a first quadrature phase component of the desired amplitude of stator current, which characterizes the flux-generating component of the stator current amplitude, in accordance with the desired amplitude of the rotor flux linkage, producing a second control input whose magnitude is proportional to the desired torque of the induction motor, in accordance with the desired and actual rotor speeds, producing a second quadrature component of the desired amplitude of stator current, which characterizes the torque-generating component of the stator current amplitude, in accordance with the magnitude of the desired torque of the induction motor, producing a static frequency of symmetrical instantaneous phase currents of the stator in steady state condition of the induction motor. The method further comprises the steps of phase-after-phase control, depending on the rotor speed and desired torque of the induction motor, of symmetrical instantaneous phase currents in the stator windings as errors between the desired and measured stator phase currents. The desired amplitude of stator current, which is equal to the desired amplitude of instantaneous stator phase currents, is in this case produced equal to the quadrature sum of the first and second quadrature components of the desired amplitude of stator current.
Besides, in this control method, the steady-state frequency of stator current is produced equal to the sum of the rotor speed and a quantity proportional to the induction motor torque. Further steps comprise producing sync sweep pulses by sweeping the steady-state frequency of stator current, the sync pulse frequency being changed proportionally to the steady-state frequency of stator current, producing asynchronous pulses whose number is proportional to the increment of the arctangent function of the desired torque of the induction motor, dicretely changing the phase of desired instantaneous stator phase currents by one increment in the forward or reverse direction as each sync pulse is supplied, said sync pulses bein used to control changes of the synchronous phase of stator current, and as each asynchronous pulse is supplied. The direction of the discrete changes of the phase of instantaneous phase currents is determined depending on the sign of the increment of the arctangent function with the arrival of an asynchronous pulse and depending on the sign of the synchronous frequency with the arrival of the sync sweep pulse. The torque-generating component of the stator current amplitude is changed in proportion to the desired torque of the induction motor. The second permanent control input is to produce a constant desired amplitude of the rotor flux linkage. The constant flux-generating component of the stator current amplitude is produced proportional to the predetermined constant value of the second control input action. The ratio of the number of asynchronous pulses to the increment of the inverse tangent function of the desired torque of the induction motor, as well as the proportionality factor of the input control action proportional to the required torque of the induction motor and the magnitude of the second component of the synchronous frequency of stator current, which is equal to the slip of the rotor flux linkage, are controlled as functions of the magnitude of the rotor resistance and mutual inductance of the induction motor assuming that the synchronous frequency of stator current is equal to the rate of rotation of the rotor flux in relation to the axis of the stator reference phase winding.
The electric drive realizing this method comprises an induction motor whose stator windings are connected, via instantaneous phase current sensors, to a pulse inverter embraced by instantaneous phase current negative feedbacks, while the rotor thereof is connected with a tachometer generator whose output is connected to one of the inputs of a proportional-plus-integral speed controller. A second input of said proportional-plus-integral speed controller is connected to an output of a speed setting unit. An output of the proportional-plus-integral speed controller is connected to inputs of three units: a stator current amplitude generator, a slip generator, and a stator current phase correction unit. The output of the slip generator and the output of the tachometer generator are connected to inputs of an adder whose output is connected to the pulse generator of the synchronous frequency of stator current. The output of the adder is also connected to an input of a unit for determining the direction of synchronous rotation of the stator current vector. The output of the stator current phase correction unit is connected to inputs of a pulse generator-counter and of a unit for determining the direction of the phase shift of the stator current vector. The outputs of a synchronous frequency generator and the pulse generator-counter are connected to the inputs of the pulse adder. The outputs of the units for determining the direction of the synchronous rotation and of the phase shift of the stator current vector are connected to inputs of a gate pulse adder. The outputs of the pulse adder and of the gate pulse adder are connected to stator current frequency control inputs of the pulse inverter. The output of stator current amplitude generator is connected to a stator current amplitude control input of the pulse inverter.
In this method the stator current vector control is synchronized in polar coordinates.
One self-contained channel is used to control the amplitude of stator current, which is equal to the length of the stator current vector.
The other self-contained channel is used to control the frequency of stator current, which is equal to the rate of rotation of the stator current vector. Non-linear dynamic stator current phase corrective action is introduced in the second control channel.
This method of induction motor control can, therefore, be referred to as a phase-frequency-current method.
The electric drive realizing this method operates as follows.
The rotor speed being zero, a constant flux-generating component of the stator current amplitude is produced in order to excite, in the induction motor, a static magnetic field characterized by a constant rotor flux in an arbitrary initial angular position which corresponds to the initial phase of stator current.
A jump in the first control input proportional to the desired torque of the induction motor produces an increment of the inverse tangent function of said first control input, which is proportional to the magnitude of the jump. The increment of the inverse tangent function results in a burst of asynchronous pulses, the number of said asynchronous pulses in the burst being equivalent to the magnitude of the increment of the inverse tangent function. Since each asynchronous pulse alters the stator current pulse by one discrete element, the stator current vector is subjected to a discrete shift in relation to the initial angular position of the rotor magnetic flux. The angle between the space vector of stator current and the space vector of the rotor flux linkage increases with each next asynchronous pulse, thus increasing the torque of the induction motor. After all asynchronous pulses are responded to, the desired induction motor torque is maintained by discrete changes of the synchronous phase of stator current as each synchronous sweep pulse arrives. The result of the simultaneous discrete turn of the stator current vector by synchronous and asynchronous pulses is synchronized control of the space vector of the rotor flux linkage, which is subjected to synchronous and cophasal angular motion concurrently with the synchronous phase assigned by the synchronous scan pulses.
In steady state conditions of the induction motor, which are characterized by constant values of induction motor torque, the rotor speed and, the amplitude of the rotor flux linkage, the rate of rotation of the space vector of the rotor flux linkage is equal to the stator current frequency. There are no asynchronous pulses, and the stator current frequency is equal to the synchronous frequency of stator current, which is assigned by synchronous scan pulses.
To summarize, the prior art method provides invariant control of the induction motor torque. It can maintain constant amplitude of the rotor magnetic flux linkage by controlling the synchronous frequency of stator current, which is equal to the rotor flux frequency, and by discrete control of the asynchronous motion of the stator current vector in relation to the rotor flux vector.
The prior art method provides for control synchronization being effected in polar coordinates by a control input action proportional to the desired motor torque and the output signal of the rate sensor without changing the reference vector of the rotor magnetic flux and employing standard transducers: a speed transducer (tachometer generator) and a current transducer.
But this prior art method and device are deficient in that their response is not sufficiently fast, the accuracy is low, and the specific torque is inadequate. The main reason of these shortcomings consists in that control is effected in accordance with the law of constant rotor flux linkage and the stator current phase control is characterized by inertia and errors, particularly when the desired torque instruction changes abruptly and asynchronous pulses arrive in sequence to follow the step. The electric drive realizing this method is deficient in that at low torque the losses for excitation of the magnetic field are too high and this narrows the motor heat margin during frequent start-stop cycles and frequent positioning of the servomechanism. At great load torque or during operational conditions, when the motor torque is twice (or thrice) as large as the rated torque value, the stator current magnitude grows substantially, power losses for heating increase, and the motor is overheated in general with respect to the minimal temperatures for a given torque. It is difficult to provide adequate dynamic accuracy of actuator motions through feedback control because of limited capabilities for momentarily boosting the motor torque. The ratio of the induction motor torque to the stator current magnitude is too low in the proposed control method based on the constant amplitude of the rotor flux linkage. This limits the safe frequency of positioning cycles during position control of robots. The efficiency of robots and other flexible manufacturing systems is seriously affected.